Three-phase three-level soft-switched pfc rectifiers

ABSTRACT

A low input-current-harmonic three-phase three-level boost rectifier includes an input stage for receiving a three-phase input voltage in relation to a neutral node and an output stage adapted to couple to at least one load. The rectifier further includes one or more switching converter stages, each having a plurality of serially-connected switches coupled to the neutral node, one of the serially-connected switches operating with a fixed duty cycle while the other of the serially-connected switches operating with a variable duty cycle, the fixed duty cycle being a substantially 50% duty cycle and the variable duty cycle being less than or equal to a substantially 50% duty cycle. The serially-connected switches are coupled to clamping diodes and clamping capacitors. The rectifier further includes one or more controllers adapted to vary the switching frequency and/or duty cycle of the plurality of switches based on at least one of a condition of the at least one load or the input voltage and includes one or more decoupling stages, each including one or more inductive elements adapted to inductively decouple the output stage from at least one of the one or more switching converter stages.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application is a continuation-in-part of copending U.S. patent application (“Copending Application”), Ser. No. 13/362,620, entitled “THREE-PHASE SOFT-SWITCHED PFC RECTIFIERS,” by Jang and Jovanović, filed on Jan. 27, 2012, and assigned the attorney docket number 36977-320233. The Copending Application is hereby incorporated by reference herein in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to front-end rectifiers with power-factor correction (PFC). In particular, the present invention relates to three-phase, three-level PFC rectifiers.

2. Discussion of the Related Art

In power converters, achieving a high efficiency in high-voltage applications is a major design challenge that requires an optimized reduction of conduction and switching losses through a careful selection of the converter topology and switching device characteristics. Specifically, a higher voltage-rated semiconductor switch exhibits larger conduction and switching losses, as compared to a counterpart with a lower voltage rating. In this context, a semiconductor switch may be any switching device, such as a MOSFET (Metal Oxide Semiconductor Field Effect Transistor), an IGBT (Insulated-Gate Bipolar Transistors), a BJT (Bipolar Junction Transistors), a SiC (Silicon-Carbide) or a GaN (Gallium-Nitride).

Generally, switching losses can be reduced and even eliminated using resonant or soft-switching techniques. However, there are only limited approaches for reducing conduction losses. In fact, once the topology and the switches with the lowest conduction losses for the required voltage rating are selected, further decrease in conduction loss is possible only by modifying the topology to utilize switches with a lower voltage rating. Multilevel converters—whose switches operate with a voltage stress that is much less than the input and output voltages—are naturally suitable for high-voltage applications.

The Copending Application describes a new, three-phase, two-switch, zero-voltage switching (ZVS), discontinuous conduction mode (DCM), PFC boost rectifier that achieves a low input-current total harmonic distortion (THD). In that PFC boost rectifier, all the switches operate under ZVS conditions, without using additional soft-switching circuitry. One implementation of the PFC rectifier of the Copending Application is shown in FIG. 1. As shown in FIG. 1, the PFC boost rectifier includes Y-connected capacitors C₁, C₂, and C₃, which create virtual neutral node N. Virtual neutral node N has the same electrical potential as the power source's neutral terminal that is not physically available for connection in a three-wire power system. Since virtual neutral node N is connected to the node between switches S₁ and S₂ and also to the node between output capacitors C_(O1) and C_(O2), the electrical potentials of these nodes are the same as the electrical potential of the neutral terminal in the balanced three-phase power source.

In addition, by connecting virtual neutral node N directly to the node between switches S₁ and S₂, decoupling of the three input currents is achieved. In such a decoupled circuit, the current in each of boost inductors L₁, L₂ and L₃ depends only on the corresponding phase voltage, which reduces the THD and increases the power factor (PF). Specifically, bridge diodes D₁-D₆ allow only the currents in phases with positive phase voltages to flow through switch S₁, when switch S₁ is turned on, and allow only the currents in phases with negative phase voltages to flow through switch S₂, when switch S₂ is on. Therefore, the boost inductor corresponding to a phase in a positive voltage half-line cycle carries positive current when switch S₁ is on, while the boost inductor corresponding to a phase in a negative voltage half-line cycle carries negative current when switch S₂ is on. During the time when switch S₁ is off, the stored energy in the boost inductor connected to the positive phase voltage is delivered to capacitor C_(R), whereas the stored energy in the boost inductor connected to the negative phase voltage is delivered to capacitor C_(R) during the time when switch S₂ is off. Because the voltage between either terminal of capacitor C_(R) and virtual neutral node N abruptly changes with a high rate (i.e., a large dV/dt value) during each switching cycle, coupled inductor L_(C) is connected between “flying” capacitor C_(R) and the output voltage V_(O) to isolate output voltage V_(O) from these fast high-voltage transitions that usually produce unacceptable common-mode electromagnetic interference (EMI) noise. As shown in FIG. 1, with coupled inductor L_(C), the node between output capacitors C_(O1) and C_(O2) can be directly connected to virtual neutral node N, which makes the output common-mode noise very low. Moreover, because of coupled inductor L_(C), parallel operations of multiple rectifiers are also possible.

To facilitate cross-reference between the figures and the detailed description, like elements are assigned like reference names or numerals.

SUMMARY

The present invention extends PFC and ZVS operations of the switches in the rectifier of FIG. 1 to three-level rectifiers that can utilize switches with a lower voltage rating and, consequently, a lower conduction loss.

According to one embodiment of the present invention, three-phase, three-level PFC rectifier topologies utilize switches with a lower voltage rating and offer improved performance over the prior art because of lower conduction losses. In one preferred embodiment, an input stage consists of three boost inductors L₁, L₂, and L₃ coupled to the three-phase input terminals and capacitors C₁, C₂, and C₃ connected in a Y or “star” configuration. Common node N of the capacitors is connected to a node between serially-connected switch pairs S₁-S₂ and S₃-S₄ and also to a node between serially connected output split capacitors C_(O1) and C_(O2). A node between serially-connected switches S₁-S₂ is connected to output capacitor C_(O1) through clamping diode D_(C1). A node between serially-connected switches S₃-S₄ is connected to output capacitor C_(O2) through clamping diode D_(C2). Capacitor C_(R), which resets the currents in the boost inductors, is connected across serially connected switches S₁-S₄ and output V_(O) by coupled or non-coupled inductor L_(C).

The rectifiers of the present invention offer a low THD in the input currents and a high PF. The rectifiers provide high efficiency power conversion using soft-switching techniques over a wide load range. This high efficiency is achieved by operating the boost inductors in DCM and by controlling the output voltages in the rectifiers using variable-frequency control. In addition, the PFC rectifiers of the present invention exhibit a reduced common-mode noise and possess automatic balancing of split capacitors C_(O1) and C_(O2), when serially-connected downstream converters are employed.

In one implementation, galvanic isolation between the input signal and the output signal is achieved using transformer TR, instead of coupled inductor L_(C). At the secondary side of transformer TR, rectifiers D_(O1) and D_(O2) and output filter including inductor L_(O) and capacitor C_(O) are coupled between the secondary winding of transformer TR and output voltage V_(O). By employing additional phase-shift or PWM control, the isolated implementations can tightly control their output voltages. Circuits may also be interleaved to increase their power handling capabilities and reduce their input or output current and voltage ripples.

The present invention is better understood upon consideration of the following detailed description and the accompanying figures.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a three-phase, two-switch, ZVS, PFC DCM boost rectifier.

FIG. 2 shows a three-phase three-level ZVS PFC DCM boost rectifier, according to an embodiment of the present invention.

FIGS. 3( a)-3(c) each show, under one of three different control schemes, gating waveforms of switches S₁-S₄ of FIG. 2 during a switching cycle, according to an embodiment of the present invention.

FIG. 4 shows a simplified model of the circuit in FIG. 2, annotated with reference directions of currents and voltages, according to an embodiment of the present invention.

FIGS. 5( a)-5(n) show topological stages illustrating voltages and currents in the model of FIG. 4 during a switching cycle, according to an embodiment of the present invention.

FIG. 6 shows the key waveforms in the model of FIG. 4, according to an embodiment of the present invention.

FIG. 7 shows a three-phase three-level ZVS PFC DCM boost rectifier with two independent loads, according to an embodiment of the present invention.

FIG. 8 shows a three-phase three-level ZVS PFC DCM boost rectifier with two independent inductors, according to an embodiment of the present invention.

FIG. 9 shows a three-phase three-level ZVS PFC DCM boost rectifier with a blocking capacitor between a virtual neutral node and the node between two split output capacitors, according to an embodiment of the present invention.

FIG. 10 shows a three-phase three-level ZVS PFC DCM boost rectifier in which snubber capacitors C_(S1), and C_(S4) are connected in parallel to switches S₁ and S₄, according to one embodiment of the present invention.

FIG. 11 shows three-phase three-level ZVS PFC DCM boost rectifiers supporting parallel or interleaved operations, according to an embodiment of the present invention.

FIG. 12 shows interleaved three-phase three-level ZVS PFC DCM boost rectifiers including DC current blocking capacitor C_(B), according to an embodiment of the present invention.

FIG. 13 shows interleaved three-phase three-level ZVS PFC DCM boost rectifiers including DC current blocking capacitors C_(B1) and C_(B2) and input filter capacitor sets C₁-C₃ and C₄-C₆, according to an embodiment of the present invention.

FIG. 14 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with two transformers and an additional phase-shift control scheme, according to an embodiment of the present invention.

FIG. 15 shows gating waveforms of switches S₁-S₈ of the circuit in FIG. 14 during a switching cycle, according to an embodiment of the present invention.

FIG. 16 shows another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifiers with two transformers and an additional phase-shift control scheme, according to an embodiment of the present invention.

FIG. 17 shows gating waveforms of switches S₁-S₈ of the circuit in FIG. 16 during a switching cycle, according to an embodiment of the present invention.

FIG. 18 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with transformer TR and split flying capacitors C_(R1) and C_(R2), according to an embodiment of the present invention.

FIG. 19 shows gating waveforms of switches S₁-S₄ in the circuit of FIG. 18 during a switching cycle, according to an embodiment of the present invention.

FIG. 20 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with a transformer and clamping diodes coupled to a node between split flying capacitors, according to an embodiment of the present invention.

FIG. 21 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with an auxiliary boost converter that uses the leakage inductances of the coupled inductor as auxiliary boost inductors, according to an embodiment of the present invention.

FIG. 22 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with an auxiliary boost converter having auxiliary boost inductors L_(H1) and L_(H2), according to an embodiment of the present invention.

FIG. 23 shows another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with an auxiliary boost converter that uses the leakage inductances of the coupled inductor as auxiliary boost inductors, according to an embodiment of the present invention.

FIG. 24 shows yet another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with an auxiliary boost converter having auxiliary boost inductors L_(H1) and L_(H2), according to an embodiment of the present invention.

FIG. 25 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with a full-bridge primary configuration and transformers TR₁ and TR₂, according to an embodiment of the present invention.

FIG. 26 shows another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with a full-bridge primary configuration and a transformer, according to an embodiment of the present invention.

FIG. 27 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with coupled two half-bridge primary configuration and series connected two transformers, according to an embodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 2 is a block diagram of a three-phase, three-level ZVS PFC DCM low input-current-harmonic boost rectifier according to an embodiment of the present invention. The input stage of the circuit in FIG. 2 includes boost inductors L₁, L₂, and L₃ coupled to capacitors C₁, C₂, and C₃ connected in a Y (“star”) configuration. The input stage of the circuit may also include an EMI filter (not shown in FIG. 2) at the three-phase input terminals. The common node N of capacitors C₁, C₂, and C₃ is connected to a node between serially-connected switch pairs S₁-S₂ and S₃-S₄ and also to a node between split output capacitors C_(O1) and C_(O2). The node between serially-connected switches S₁-S₂ is connected to output capacitor C_(O1) through clamping diode D_(C1), so that the voltage across switch S₂ is clamped to the voltage across capacitor C_(O1), which is preferably one half of output voltage V_(O). The node between serially-connected switches S₃-S₄ is connected to output capacitor C_(O2) through clamping diode D_(C2) and the voltage across switch S₃ is also clamped to one-half of output voltage V_(O), preferably. Clamping capacitor C_(C) is connected across split output capacitors C_(O1) and C_(O2), and is pre-charged to its steady-state average voltage of one-half of the output voltage through the loop consisting of capacitor C_(O2), the body diode of switch S₂, pre-charge resistor R_(PR), and a winding of coupled inductor L_(C). Capacitor C_(R) which resets the boost inductor currents is connected across serially-connected pairs of switches S₁-S₂ and S₃-S₄, and is decoupled from output voltage V_(O) by inductor L_(C). The average voltage across capacitor C_(R) is equal to output voltage V_(O) since the average voltage across the windings of L_(C) is zero. The voltages across switches S₁ and S₄ are clamped to the difference in voltage across capacitor C_(R) and C_(C) by the body diodes of switch S₄ and switch S₁. Since this difference in voltage is equal to one-half of output voltage V_(O), the voltage across each of the four switches S₁-S₄ in the circuit in FIG. 2 is one-half of output voltage V_(O). Generally, any kind of switch that is capable of conducting current in both directions and blocking voltage in one direction with an antiparallel diode (e.g., a MOSFET or an IGBT) is suitable for this application.

The Y-connected capacitors C₁, C₂, and C₃ create virtual ground node N, i.e., a node with the same electrical potential as the input (source) voltage neutral wire that is not physically available or connected in a three-wire power system. By connecting virtual neutral node N directly to the nodes between switch pairs S₁-S₂ and S₃-S₄, decoupling of the three input currents is achieved. In such a decoupled circuit, the current in each of boost inductors L₁, L₂ and L₃ depends only on the corresponding phase voltage, which reduces the THD and increases the PF. Specifically, in the circuit in FIG. 2, bridge diodes D₁-D₆ allow only a positive input voltage to deliver currents through series connected switches S₁ and S₂ when these switches are turned on and a negative input voltage to deliver currents through series connected switches S₃ and S₄ when these switches are turned on. The switches may each be turned on at a substantially zero voltage across the switches. Therefore, any boost inductor in a phase with a positive voltage half-line cycle carries positive current when switches S₁ and S₂ are on, while any boost inductor in a phase with a negative voltage half-line cycle carries negative current when switches S₃ and S₄ are turned on. During the time when switches S₁ and S₂ are turned off, the stored energy in the boost inductor connected to the positive phase voltage is delivered to capacitor C_(R), while the stored energy in the boost inductor connected to the negative phase voltage is delivered to capacitor C_(R) during the time when switches S₃ and S₄ are turned off.

Because in every switching cycle the voltage across capacitor C_(R) changes rapidly (i.e., with a large dV/dt value), coupled inductor L_(C) connects between “flying” capacitor C_(R) and output voltage V_(O) to isolate output voltage V_(O) from these fast high-voltage transitions that usually produce unacceptable common-mode EMI noise. With coupled inductor L_(C), the output common-mode noise is very low, as the noise is confined to the S₁-S₂-S₃-S₄-C_(R) loop. Moreover, because coupled inductor L_(C) provides impedance between output voltage V_(O) and switches S₁-S₄, parallel or interleaving operations of multiple rectifiers are possible.

To achieve a low input-current THD, high PF, and soft-switching of switches S₁-S₄ over a wise load range, the circuit in FIG. 2 must operate in DCM with a low-bandwidth output-voltage control scheme. This control scheme can be implemented in a variety of ways. FIG. 3( a) shows gating waveforms for controlling switches S₁-S₄ in FIG. 2 under a variable switching-frequency control scheme. Under the variable-frequency control scheme of FIG. 3( a), switch pairs S₁-S₂ and S₃-S₄ are switched in a complementary fashion, with a small dead time t_(d) between their commutation instants to enable the switch pair that is about to turn on to achieve ZVS. Since dead time t_(d) is very small in comparison with switching period T_(S), the effect of the dead time on the duty cycle is negligible, i.e., the duty cycle of each switch pair is approximately 50%.

FIG. 3( b) shows a second control scheme for switches S₁-S₄ of FIG. 2. Under this second control scheme, switches S₁ and S₄ are switched at a constant frequency in a complementary fashion with a small dead time in-between, i.e., each switch operates at an approximate 50% duty cycle. Switches S₂ and S₃, whose turn-on instants are synchronized with the turn on instants of S₁ and S₄, respectively, are each pulse-width modulated to provide regulation of output voltage V_(O).

FIG. 3( c) shows a third alternative control scheme for switches S₁-S₄ of FIG. 2. Under this third control scheme, in each of switch pairs S₁-S₄ and S₂-S₃, the switches within the switch pair are switched at a constant frequency in a complementary fashion with a fixed duty cycle of approximately 50%. The control scheme provides a phase shift between the switching instants of the S₁-S₄ pair and the corresponding switching instants of the S₂-S₃ pair. In this phase-shift control scheme, the output voltage is zero when the phase shift is zero and reaches a maximum when the phase shift is 180⁰ (i.e., when the phase shift is T_(S)/2).

The control schemes illustrated in FIGS. 3( a)-(c) may be used in a combination. Namely, the variable switching frequency control scheme of FIG. 3( a) can be used in any combination with the constant frequency PWM control scheme of FIG. 3( b), or with the phase-shift control scheme of FIG. 3( c), to limit the switching frequency range. For example, in the output voltage-regulated converter of FIG. 2, the switching frequency increases as the load decreases. Thus, the frequency range can be reduced by a variable switching-frequency control scheme at full and medium loads, while switching over to a constant frequency control scheme at lighter loads.

FIG. 4 shows a simplified model of the circuit of FIG. 2, annotated with reference directions of currents and voltages, according to an embodiment of the present invention. To simplify analysis, ripple voltages of the input and output filter capacitors in FIG. 2 (i.e., capacitors C₁, C₂, C₃, C_(O1) and C_(O2)) are considered negligible in this model, so that the voltage across the input and output filter capacitors can be represented by constant-voltage source V_(AN), V_(BN), V_(CN), V_(O1), and V_(O2). Also, in the on state, the semiconductor switches exhibit zero resistance (i.e., they are short circuits). However, the output capacitances of the switches are not neglected in this model. Coupled inductor L_(C) in FIG. 2 is modeled as a two-winding ideal transformer with magnetizing inductance L_(M) and leakage inductances L_(LK1) and L_(LK2). In this model, the average voltage across flying capacitor C_(R) is substantially equal to output voltage V_(O)=V_(O1)+V_(O2) and the average voltage across clamping capacitor C_(C) is substantially equal to one half of output voltage V_(O). The reference directions of currents and voltages in FIG. 4 correspond to a 60-degree segment of the line cycle (i.e., when V_(AN)>0, V_(BN)<0, and V_(CN)<0).

FIGS. 5( a)-5(n) show topological stages illustrating voltages and currents in the model of FIG. 4 during a switching cycle, according to an embodiment of the present invention. FIG. 6 shows the key waveforms in the model of FIG. 4, according to an embodiment of the present invention.

The waveforms of the gating signals of switches S₁-S₄ in FIG. 6 show that the control scheme used combines a variable-frequency control scheme and a constant-frequency phase-shift control scheme. In FIG. 6, switches S₁ and S₄ operate in a complementary manner. Likewise, switches S₂ and S₃ operate in a complementary manner. In either case, a short dead time is provided between the commutation instants (i.e., each switch operates with a fixed duty cycle of approximately 50%). This gating strategy enable ZVS in the switches that are about to turn on. To regulate output voltage V_(O) in the presence of input voltage and output load variations, a variable switching frequency control scheme is employed. However, to limit the control-frequency range and the consequential switching losses, the variable-frequency control scheme is assisted by a phase-shift control scheme at light loads or high input voltage, or both.

In the model of FIG. 4, the minimum frequency occurs when both a full load and the minimum input voltage are present, while duty cycle is set substantially at 50%. The maximum frequency occurs when a light load and maximum input voltage are present, while duty cycle is set below 50%. If necessary, the rectifier of the present invention can operate in a controlled burst mode or pulse skip mode at no load or at a very light load, to avoid operation at an unnecessarily high switching frequency. Other control strategies could also be applied to this circuit, including constant-frequency PWM control and phase-shift control as shown in FIGS. 3( b) and 3(c) discussed above.

Referring to FIGS. 5( a) and 6, before switch S₂ is turned off at t=T₁, inductor current i_(L1) flows through closed switches S₁ and S₂. The slope of inductor current i_(L1) is equal to V_(AN)/L₁ and the peak of the inductor current at t=T₁ is approximately

$\begin{matrix} {{I_{L\; 1{({PK})}} = {\frac{V_{AN}}{L_{1}} \times {DT}_{S}}},} & (1) \end{matrix}$

where V_(AN) is line-to-neutral voltage and DT_(S) is the portion of the switching period T_(S) during which switches S₁ and S₂ are both closed (i.e., D is the effective duty cycle). Because the dead time between the turning-off of switch S₁ and the turning-on of switch S₄ is very short relative to switching period T_(S), the effect of the dead time is neglected in Equation (1). During the period between times T_(O) and T₁, current i_(O1) decreases at a rate of −V_(O1)/(L_(M)+L_(LK1)) while current i_(O2) increases at a rate of (V_(CR)−V_(O1))/(L_(M)+L_(LK2)). Magnetizing current i_(M) is the difference between currents i_(O1) and i_(O2).

The magnetizing inductance value of coupled inductor L_(M) is selected to be sufficiently large, such that its ripple current does not significantly affect rectifier operation. As shown in FIG. 2, the windings of inductor L_(C) are coupled in such a way as to cancel the magnetic fluxes from the differential current of the two windings, so that the large magnetizing inductance can be achieved by a small gap in the core without saturation. Since the ripples in currents i_(O1) and i_(O2) are considered negligible in this model, further discussion is omitted, although the ripple currents are still shown in the topological stages in FIG. 5.

At t=T₁, switch S₂ turns off, inductor current i_(L1) begins to charge the parasitic output capacitance of switch S₂ (FIG. 5( b)). As the sum of the voltages across switches S₂ and S₃ is clamped to clamping capacitor voltage V_(CC), the output capacitance of switch S₃ discharges at the same rate as the output capacitance of switch S₂ is being charged until the output capacitance of switch S₂ is fully charged and clamping diode D_(C1) starts to conduct at t=T₂, as shown in FIG. 5( c) and FIG. 6. Shortly after t=T₂, switch S₃ turns on under ZVS condition.

During interval [T₂, T₃], because clamping diode D_(C1) is forward biased, inductor current i_(L1) linearly decreases. The slope of inductor current i_(L1) is equal to (V_(AN)−V_(O1))/L₁ and the inductor current at t=T₃ is approximately

$\begin{matrix} {{\left. I_{L\; 1} \right|_{t = {T\; 3}} = {\frac{V_{AN} - {\left( {1 - {2\; D}} \right)V_{O\; 1}}}{2\; L_{1}} \times T_{S}}},} & (2) \end{matrix}$

At t=T₃, when switch S₁ turns off, inductor current i_(L1) begins charging the output capacitance of switch S₁, as shown in FIG. 5( d). Because the sum of the voltages across switches S₁ and S₄ is clamped to the voltage difference between flying capacitor voltage V_(CR) and clamping capacitor voltage V_(CC), the output capacitance of switch S₄ discharges at the same rate as the output capacitance of switch S₁ is charged, until the output capacitance of switch S₄ is fully discharged and the anti-parallel body diode of switch S₄ starts to conduct at t=T₄, as shown in FIG. 5( e) and FIG. 6. At t=T₅, switch S₄ turns on a ZVS condition and inductor current i_(L1) is commutated from the antiparallel body diode of switch S₄ to the switch, as illustrated in FIG. 5( f). Because the body diode of switch S₄ is forward biased and switch S₃ is switched on, inductor currents i_(L2) and 4₃ begin to linearly increase after t=T₄. At t=T₅, switch S₄ turns on under a ZVS condition and inductor currents i_(L2) and i_(L3) are commutated from the antiparallel body diode of switch S₄ to the switch, as illustrated in FIG. 5( f), until inductor current i_(L1) decreases to zero at t=T₆. To maintain DCM operation, the minimum voltage V_(CR (MIN)) across “flying” capacitor C_(R), which is equal to output voltage V_(O), is provided by:

$\begin{matrix} {V_{{CR}{({MIN})}} = {\frac{V_{{AN}{({PK})}}}{1 - D} = {\frac{\sqrt{2}}{\sqrt{3}\left( {1 - D} \right)} \times V_{{L - L},{RMS}}}}} & (3) \end{matrix}$

where V_(AN-PK) is the peak line-to-neutral voltage.

During the T₄-T₆ interval, because inductor currents i_(L2) and i_(L3) both flow in an opposite direction from inductor current i_(L1), the average current through switches S₃ and S₄ is reduced, resulting in reduced power losses in the switches.

During the period between t=T₆ and t=T₇, inductor currents i_(L2) and i_(L3) continue to flow through switches S₃ and S₄, as illustrated in FIG. 5( g). As shown in FIG. 6, the slopes of inductor currents i_(L2) and i_(L3) during this period are equal to −V_(BN)/L₂ and −V_(CN)/L₃, respectively. The peaks of the inductor currents at the moment when switch S₃ turns off at t=T₇ are approximately

$\begin{matrix} {I_{L\; 2{({PK})}} = {{- \frac{V_{BN}}{L_{2}}} \times {DT}_{S}\mspace{14mu} {and}}} & (3) \\ {I_{L\; 3{({PK})}} = {{- \frac{V_{CN}}{L_{3}}} \times {{DT}_{S}.}}} & (4) \end{matrix}$

Therefore, as seen from Equations (1), (3), and (4), the peak inductor current is proportional to its corresponding input voltage, as long as duty cycle D is substantially constant during one half of the line cycle.

After switch S₃ turns off at t=T₇, inductor currents i_(L2) and i_(L3) start to simultaneously charge the output capacitance of switch S₃ and discharge the output capacitance of switch S₂, as shown in FIG. 5( h), until t=T₈, when the output capacitance of switch S₃ is fully charged and clamping diode D_(C2) starts to conduct at t=T₈, as shown in FIG. 5( i) and FIG. 6. After t=T₈, switch S₂ turns on under a ZVS condition.

At time t=T₈, because clamping diode D_(C2) is forward biased, inductor currents i_(L2) and i_(L3) begin to linearly increase until inductor current i_(L3) reaches zero at time t=T₉. The slopes of inductor currents i_(L2) and i_(L3) are equal to (−V_(BN)+V_(O2))/L₂ and (−V_(CN)+V_(O2))/L₃, respectively. Inductor current i_(L2) at t=T₁₀, when switch S₄ turns off, is approximately

$\begin{matrix} {\left. I_{L\; 2} \right|_{t = {T\; 10}} = {\frac{{- V_{BN}} + {\left( {1 - {2\; D}} \right)V_{O\; 2}}}{2\; L_{2}} \times {T_{S}.}}} & (5) \end{matrix}$

At t=T₁₀, when switch S₄ turns off, inductor current i_(L2) starts to charge the output capacitance of switch S₄, as shown in FIG. 5( k). Because the sum of the voltages across switches S₁ and S₄ is clamped to the voltage difference between flying capacitor voltage V_(CR) and clamping capacitor voltage V_(CC), the output capacitance of switch S₁ discharges at the same rate as the output capacitance of switch S₄ is being charged, until the output capacitance of switch S₁ is fully discharged and the anti-parallel body diode of switch S₁ starts to conduct at t=T₁₁, as shown in FIG. 5( l) and FIG. 6. At t=T₁₂, switch S₁ turns on under a ZVS condition and inductor currents i_(L2) is commutated from the antiparallel body diode of switch S₁ to the switch itself, as shown in FIG. 5( m). At this time, because switches S₁ and S₂ are both on, inductor current i_(L1) begins to linearly increase after t=T₁₁. During period T₁₂-T₁₃. increasing inductor current i_(L2), continues to flow through switches S₁ and S₂, as shown in FIG. 5( m). Finally, after inductor current i_(L2) reaches zero at t=T₁₃, a new switching cycle begins, as shown in FIG. 5( n).

The harmonic content of the average inductor currents i_(L1)-i_(L3) shown in FIG. 6 is dominated by the 3^(rd) harmonic. However, as the neutral wire in a three-wire power system is not available (or not connected), the phase currents cannot contain the triplen harmonics (the 3^(rd) harmonic and the odd multiples of the 3rd harmonic). As a result, the circuit of the present invention exhibits a very low THD and a high PF, as the remaining harmonics contribute less than 1-2% of total current distortion.

A PFC rectifier of the present invention may be implemented in many ways. For example, FIG. 7 shows an implementation supporting independent loads R₁ and R₂. Since the two-switch rectifier automatically balances the voltages across output capacitors C_(O1) and C_(O2), no additional voltage-balancing circuit is required. Natural voltage-balancing is achieved because in the circuit in FIG. 7, the average voltages across switch pairs _(S) ₁-S₂ and S₃-S₄ are equal to voltages V_(O1) and V_(O2) across capacitors C_(O1) and C_(O2), respectively, as the average voltages across the windings of inductor L_(C) are zero. The average voltages of switch pairs S₁-S₂ and S₃-S₄ are equal to V_(CR)/2, so that V_(O1)=V_(O2)=V_(CR)/2.

Alternatively, the circuits of the present invention may also be implemented using independent inductors L_(C1) and L_(C2), as shown in FIG. 8, or with blocking capacitor C_(B), as shown in FIG. 9. By coupling blocking capacitor C_(B) between virtual neutral node N and output capacitors C_(O1) and C_(O2) (FIG. 9), low frequency currents circulating between virtual neutral node N, switches S₂ and S₃ and the common node between capacitors C_(O1) and C_(O2) can be reduced.

FIG. 10 shows an embodiment in which snubber capacitors C_(S1) and C_(S4) are connected in parallel with switches S₁ and S₄. By adding snubber capacitors C_(S1) and C_(S4), turn-off losses in switches S₁ and S₄ can be significantly reduced.

FIG. 11 shows a circuit of the present invention that combines two sub-circuits (“PFC stages”) which operate in parallel or are interleaved, in accordance with one embodiment of the present invention. The PFC stages in FIG. 11 each include an input portion of the circuit similar to that shown in FIG. 2 (from the three-phase input voltage up to coupled inductor L_(C)). In parallel operation, switches S₁, S₂, S₅, and S₆ are driven by a first gating signal, while switches S₃, S₄, S₇, and S₈ are driven by a second gating signal that is complementary the first gating signal. In an interleaved operation, switches S₁, S₂, S₇, and S₈ are driven by the first gating signal, while switches S₃, S₄, S₅, and S₆ are driven by the second gating signal. The parallel and interleaved operations are made possible by coupled inductors L_(C1) and L_(C2), which provide decoupling impedances between the output portion and the switches.

FIG. 12 shows an interleaved three-phase three-level ZVS PFC DCM boost rectifier including DC current blocking capacitor C_(B), according to one embodiment of the present invention. By including blocking capacitor C_(B), a DC current circulating between the common nodes of switches S₂, S₃, S₆, and S₇ and the common node of output capacitors C_(O1) and C_(O2) can be prevented.

FIG. 13 shows an interleaved three-phase three-level ZVS PFC DCM boost rectifier including DC current blocking capacitors C_(B1) and C_(B2) and input filter capacitor sets C₁-C₃ and C₄-C₆, according to one embodiment of the present invention.

FIG. 14 shows three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier including transformers TR, and TR₂ and an additional phase-shift control scheme. A tightly controlled output voltage can be achieved without additional switches, when two isolated PFC rectifiers are connected in parallel. As shown in FIG. 14, transformers TR₁ and TR₂ replace coupled inductors L_(C1) and L_(C2) of FIG. 13. On the secondary side of transformers TR₁ and TR₂, rectifiers D_(O1)-D_(O4) and the output filter formed by inductor L_(O) and capacitor C_(O) are coupled between the serially-connected secondary windings of transformers TR₁ and TR₂ and output voltage V_(O). Furthermore, using an additional phase-shift or PWM control scheme, the isolated circuits connected to the primary windings of transformers TR₁ and TR₂ can tightly control the output voltages to minimize unnecessary voltage ripples.

FIG. 15 shows gating waveforms of switches S₁-S₈ for the circuit of FIG. 14. Switches S₁ and S₂ of the first PFC stage turn on before switches S₇ and S₈ of the second PFC stage turn off. This phase shift between the two set of switches creates PWM voltage waveforms V_(P1) and V_(P2) across the primary windings of transformers TR₁ and TR₂, as shown in FIG. 15. Switches S₂, S₃, S₆, and S₇ turn off slightly earlier than switches S₁, S₄, S₅, and S₈ turn off, which ensures that the switch voltages be clamped to one-half of the output voltage V_(O). Switches S₁-S₈ operate with a slowly varying switching frequency and near 50% duty cycle to achieve high PF and low THD, while the phase shift is used to tightly regulate the output voltage.

FIG. 16 shows another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with two transformers and an additional phase-shift control scheme. The circuit of FIG. 16 is implemented using clamping diodes D_(C1) and D_(C2) that connect both the node between switches S₁ and S₂ and the node between switches S₃ and S₄ to the common node between split flying capacitors C_(R1) and C_(R2), so as to clamp the voltages across switches S₁ and S₄, respectively. The circuit of FIG. 16 is also implemented with the secondary windings of transformers TR₁ and TR₂ connected in series to provide current sharing.

FIG. 17 shows gating waveforms of switches S₁-S₈ for the circuit in FIG. 16. Switches S₁ and S₂ of the first PFC stage turn on before switches S₇ and S₈ of the second PFC stage turn off. The phase shift between the two switching instants creates PWM voltage waveforms V_(P1) and V_(P2) across the primary windings of transformers TR₁ and TR₂, as shown in FIG. 17. Switches S₁, S₄, S₅, and S₈ turn off slightly earlier than switches S₂, S₃, S₆, and S₇ turn off, which ensures that the switch voltages are clamped to one-half of output voltage V_(O).

Generally, most off-line power supplies of today have two cascaded stages—a front-end PFC rectifier stage and an output isolated dc-dc converter stage. As a result, although the output voltage of the front-end PFC rectifier contains rectified line frequency voltage ripples, the secondary isolated dc-dc converter stage can regulate its output voltage by its own high-frequency bandwidth feedback control. A single-stage approach combines the functions of the two cascaded stages. As a result, a single-stage rectifier should achieve PFC as well as tight regulation of the output voltage.

FIG. 18 shows a single-stage rectifier with an isolated output. The primary side of the isolated circuit of FIG. 18 is similar to the circuit of FIG. 2, except that the circuit in FIG. 18 includes flying capacitors C_(R1) and C_(R2) and additional transformer TR that has a primary winding and a center-tap secondary winding. On the secondary side of transformer TR, rectifiers D_(O1) and D_(O2) and output filter components L_(O) and C_(O) are coupled between the secondary winding of transformer TR and output voltage V_(O). To achieve high PF and low THD, the switching frequency of switches S₁-S₄ are kept nearly constant during a half cycle of the line frequency. To keep the switching frequency substantially constant, a low-pass filter may be included in the path of the frequency-control feedback loop. As a result, the bandwidth of the frequency control feedback loop is lower than the line frequency, which produces significant voltage ripples across flying capacitor C_(R1) and C_(R2) at rectified line frequency (e.g., six times higher frequency than the fundamental frequency of the line). However, a tight regulation of the output voltage is achieved by an additional a high-pass filter in the path of the phase-shift or PWM high-band-width control feedback loop.

FIG. 19 shows gating waveforms of switches S₁-S₄ for the circuit of FIG. 18. The gating signals of switches S₁ and S₄ are phase shifted with respect to the corresponding gating signals of switches S₂ and S₃. This phase shift creates PWM voltage waveforms across the primary windings of transformer TR. Switches S₁-S₄ operate with a slowly varying switching frequency and near 50% duty cycle to achieve high PF and low THD, while the variation of the phase shift is used to tightly regulate the output voltage.

FIG. 20 shows another three-phase, three-level single-stage isolated ZVS PFC DCM boost rectifier with a transformer and an additional phase-shift control scheme. The circuit is implemented by connecting clamping diodes D_(C1) and D_(C2) to the node between split flying capacitors C_(R1) and C_(R2), so as to clamp the voltages across switches S₁ and S₄, respectively.

FIG. 21 shows yet another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with an auxiliary boost converter that consists of switch S_(H) and diodes D_(H1) and D_(H2). The leakage inductances of coupled inductor L_(C) serve as a boost inductor for the auxiliary boost converter. The auxiliary boost converter begin operating immediately after the tree-phase input voltage goes to zero, to extend the output voltage regulation time.

FIG. 22 shows another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with an auxiliary boost converter that consists of switch S_(H), diodes D_(H1) and D_(H2), and auxiliary boost inductors L_(H1) and L_(H2).

FIG. 23 shows yet another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with an auxiliary boost converter that consists of switch S_(H), diodes D_(H1)-D_(H3), and auxiliary boost inductors L_(H1) and L_(H2). In FIG. 23, series diode D_(H1) of FIG. 22 that connects between input bridge diodes D₁-D₃ and switch S₁ has been eliminated. This approach is desirable for applications that require high efficiency operation. The leakage inductances of coupled inductor L_(C) serve as a boost inductor of the auxiliary boost converter.

FIG. 24 shows another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with an auxiliary boost converter that consists of switch S_(H), diodes D_(H1)-D_(H4), and auxiliary boost inductors L_(H1) and L_(H2).

FIG. 25 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with switches S₁-S₈ and an isolated output, according to one embodiment of the present invention. The primary side of the isolated circuit in FIG. 25 is similar to the circuit of FIG. 2, except that the circuit has split flying capacitors C_(R1) and C_(R2), auxiliary switches S₅-S₈, additional clamping diodes D_(C3) and D_(C4), and series connected transformers TR₁ and TR₂ each consisting of a primary winding and a center-tap secondary winding. On the secondary side of transformers TR₁ and TR₂, rectifiers D_(O1)-D_(O4) and output filter components L_(O1), L_(O2) and C_(O) are coupled between the secondary windings of transformers TR₁ and TR₂ and the output. Blocking capacitor C_(B) is connected in series with transformers TR₁ and TR₂ to eliminate any DC current through the primary windings of the transformers. The controller operates main switches S₁-S₄ and auxiliary switches S₅-S₈ to achieve frequency and phase-shift control, using the gating signal waveforms in FIGS. 15 and 17. The isolated single-stage implementation can achieve a tight control of its output voltage in addition to high PF and low THD.

FIG. 26 shows another three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with switches S₁-S₈ and an isolated output, according to an embodiment of the present invention. The primary side of the isolated circuit in FIG. 26 is similar to the circuit of FIG. 2, except for split flying capacitors C_(R1) and C_(R2), auxiliary switches S₅-S₈, additional clamping diodes D_(C3) and D_(C4), and transformer TR that consists of a primary winding and a center-tap secondary winding. On the secondary side of transformer TR, rectifiers D_(O1)-D_(O2) and output filter components L_(O) and C_(O) are coupled between the secondary windings of transformer TR and output voltage V_(O). Blocking capacitor C_(B) is connected in series with transformer TR to eliminate any dc current flowing through the primary windings of transformer TR.

FIG. 27 shows a three-phase three-level single-stage isolated ZVS PFC DCM boost rectifier with coupled half-bridge primary configurations and series connected transformers TR₁ and TR₂, according to one embodiment of the present invention.

Although the isolated circuits shown in FIGS. 14-27 show the output circuit with a center-tap secondary winding of the transformer and two output diodes, the output circuit may also be implemented using any known output rectifier topology, such as the current doubler rectifier, the full-bridge rectifier, the half-bridge with voltage doubler capacitors, within the scope of the present invention. In addition, a synchronous rectifier can be employed instead of low-voltage diode rectifiers. 

1. A low input-current-harmonic three-phase boost rectifier, comprising: an input stage for receiving a three-phase input voltage relative to a neutral node, the input stage comprising three terminals, each terminal being coupled to one of the three phases of the three-phase input voltage; an output stage adapted to couple to at least one load, the output stage comprising a plurality of serially connected output capacitors that are coupled between the neutral node and said load; one or more input capacitors each coupled between a terminal of said input stage and said neutral node; one or more switching converter stages, each comprising: two or more pluralities of serially-connected switches, each plurality of switches being coupled to said neutral node, at least one of the switches of said plurality of serially-connected switches operating at a variable frequency with a substantially 50% duty cycle and at least one of the switches of said plurality of serially-connected switches operating with a variable duty cycle that is less than or substantially equal to 50%; one or more clamping capacitors each coupled to a common node of the switches of said pluralities of serially-connected switches; one or more clamping diodes each clamping one of the common nodes to one of the voltages across the serially connected output capacitors; one or more boost inductors each coupled to one of the terminals of the input stage to receive a corresponding one of the phases of the three-phase input voltage; one or more input rectifier diodes each coupled between one of the boost inductors and a terminal of one of said pluralities of serially-connected switches; and one or more flying capacitors coupled across said pluralities of serially-connected switches; and one or more controllers adapted to vary at least one of (a) the switching frequency and (b)/the duty cycle, based on at least one of: conditions at the load, the input voltage, and the output voltage; and one or more decoupling stages, each comprising one or more inductive elements adapted to decouple the output stage from at least one of the one or more switching converter stages.
 2. The low input-current-harmonic boost rectifier of claim 1, wherein the maximum voltage across each switch is substantially one half of the output voltage.
 3. The low input-current-harmonic boost rectifier of claim 1, wherein one or more switches of the plurality of switches are turned on at a substantially zero voltage switching condition.
 4. The low input-current-harmonic boost rectifier of claim 1, wherein the variable switching frequency varies between (a) a minimum frequency determined by a full load and a minimum input voltage, and (b) a maximum frequency determined by a light load and a maximum input voltage.
 5. The low input-current-harmonic boost rectifier of claim 1, wherein the variable duty cycle varies between (a) a maximum duty cycle determined by a minimum input voltage, and (b) a minimum duty cycle determined by a maximum input voltage.
 6. The low input-current-harmonic boost rectifier of claim 1, wherein the variable duty cycle varies between (a) a maximum duty cycle determined by a full load and a minimum input voltage, and (b) a minimum duty cycle determined by a light load and a maximum input voltage.
 7. The low input-current-harmonic boost rectifier of claim 1, wherein the variable duty cycle is varied based on a pulse-width modulation control scheme.
 8. The low input-current-harmonic boost rectifier of claim 1, wherein the variable duty cycle is varied based on a phase shift control scheme.
 9. The low input-current-harmonic boost rectifier of claim 1, wherein the controllers are adapted to operate in a burst mode over a range between a light load and no load.
 10. The low input-current-harmonic boost rectifier of claim 1, wherein the inductive elements are magnetically coupled.
 11. The low input-current-harmonic boost rectifier of claim 1, wherein each one of the output capacitors is adapted to couple across at least one load.
 12. The low input-current-harmonic boost rectifier of claim 1, wherein the voltage across each one of the output capacitors are substantially equal.
 13. The low input-current-harmonic boost rectifier of claim 1, further comprising a blocking capacitor coupled between the neutral node and one of the output capacitors of the output stage.
 14. The low input-current-harmonic boost rectifier of claim 1, wherein the output capacitors of the output stages are coupled to an input neutral wire of a three-phase four-wire input system.
 15. The low input-current-harmonic boost rectifier of claim 1, wherein one or more pre-charge resistors are coupled between one of the clamping capacitors and one of the flying capacitors.
 16. The low input-current-harmonic boost rectifier of claim 1, wherein one or more inrush current control circuits are coupled between the input voltage and the capacitors.
 17. The low input-current-harmonic boost rectifier of claim 1, wherein the input stage further comprises input electromagnetic interference (EMI) filters.
 18. The low input-current-harmonic boost rectifier of claim 1, wherein the input bridge diodes are synchronous rectifiers.
 19. The low input-current-harmonic boost rectifier of claim 1, wherein each switch comprises a metal-oxide-semiconductor field-effect transistor (MOSFET) or an insulated gate bipolar transistor (IGBT) with an antiparallel diode.
 20. The low input-current-harmonic boost rectifier of claim 1, wherein each switch is connected to at least one snubber capacitor in parallel.
 21. The low input-current-harmonic boost rectifier of claim 1, wherein the switching converter stages are each coupled to a corresponding one of the one or more decoupling stages, and wherein the controllers are further adapted to operate the switches in an interleaved or a parallel manner.
 22. The low input-current-harmonic boost rectifier of claim 1, wherein the switching converter stages are each being coupled to a corresponding one of the one or more decoupling stages through one or more blocking capacitors, and wherein the controllers are further adapted to operate the switches in an interleaved or a parallel manner.
 23. A low input-current-harmonic three-phase boost rectifier, comprising: an input stage for receiving a three-phase input voltage relative to a neutral node, the input stage comprising three terminals, each terminal being coupled to one of the three phases of the three-phase input voltage; an output stage adapted to couple to at least one load, the output stage comprising a plurality of serially connected output capacitors that are coupled to between the neutral node and said load; one or more input capacitors each coupled between a terminal of the input stage and said neutral node; one or more switching converter stages, each comprising: two or more pluralities of serially-connected switches that are coupled to said neutral node, at least one of said serially-connected switches operating at a variable frequency with a substantially 50% duty cycle and at least one of said serially-connected switches operating with a variable duty cycle that is less than or substantially equal to 50%; one or more clamping capacitors each being coupled across common nodes between switches of said pluralities of serially-connected switches; one or more clamping diodes each clamping one of the common nodes to one of the voltages across the serially connected output capacitors; one or more boost inductors that are each coupled to one of the terminals of the input stage; one or more input rectifier diodes each coupled between one of the boost inductors and a terminal of one of said serially-connected switches; and one or more flying capacitors coupled across said switches; and one or more controllers adapted to vary the switching frequency and the duty cycle, based on at least one of: conditions of the load, the input voltage, and the output voltage; and one or more isolation stages, each comprising one or more transformers adapted to isolate the output stage from at least one of the switching converter stages.
 24. The low-input-harmonic boost rectifier of claim 23, further comprising one or more decoupling stages, each decoupling stage comprising one or more inductive elements adapted to decouple a plurality of energy storage capacitors that are coupled to the neutral node.
 25. The low input-current-harmonic boost rectifier of claim 23, wherein said at least one transformer comprises at least one primary winding and one secondary winding, said secondary winding being coupled to a plurality of rectifiers.
 26. The low input-current-harmonic boost rectifier of claim 23, wherein said flying capacitors are coupled across said switches and are also coupled to at least one transformer to form at least one half-bridge converter.
 27. The low input-current-harmonic boost rectifier of claim 25, wherein said secondary winding and said rectifiers are adapted to form a full-wave rectifier.
 28. The low input-current-harmonic boost rectifier of claim 25, wherein said secondary winding further comprising a center-tapped winding, and wherein said center-tapped winding and said rectifiers are adapted to form a full-wave rectifier.
 29. The low input-current-harmonic boost rectifier of claim 25, wherein said secondary winding and said rectifiers are adapted to form a current-doubler rectifier.
 30. The low input-current-harmonic boost rectifier of claim 25, wherein said secondary-side rectifiers comprise synchronous rectifiers.
 31. The low input-current-harmonic boost rectifier of claim 25, further comprising one or more output filters on the secondary side of the at least one transformer.
 32. The low input-current-harmonic boost rectifier of claim 23, wherein one or more of the serially-connected switches are turned on under a substantially zero voltage switching condition.
 33. The low input-current-harmonic boost rectifier of claim 24, further comprising one or more hold-up time switches coupled across the inductive elements and one or more diodes coupled between the input rectifier diodes and the plurality of serially-connected switches.
 34. The low input-current-harmonic boost rectifier of claim 33, wherein the one or more hold-up time switches operates just after the input voltage drop out to extend the hold-up time.
 35. The low input-current-harmonic boost rectifier of claim 24, further comprising one or more hold-up time switches coupled across the inductive elements, one or more charging diodes coupled between the boost inductors and the hold-up time switches, and one or more diodes coupled between the inductive elements and the plurality of serially-connected switches.
 36. The low input-current-harmonic boost rectifier of claim 23, wherein the variable switching frequency varies between (a) a minimum frequency determined by a full load and a minimum input voltage, and (b) a maximum frequency determined by a light load and a maximum input voltage.
 37. The low input-current-harmonic boost rectifier of claim 23, wherein the variable duty cycle varies between (a) a maximum duty cycle determined by a minimum input voltage, and (b) a minimum duty cycle determined by a maximum input voltage.
 38. The low input-current-harmonic boost rectifier of claim 23, wherein the variable duty cycle varies between (a) a maximum duty cycle determined by a full load and a minimum input voltage, and (b) a minimum duty cycle determined by a light load and a maximum input voltage.
 39. The low input-current-harmonic boost rectifier of claim 23, wherein the variable duty cycle is varied based on a pulse-width modulation control scheme.
 40. The low input-current-harmonic boost rectifier of claim 23, wherein the variable duty cycle is varied based on a phase shift control scheme.
 41. The low input-current-harmonic boost rectifier of claim 23, wherein the controllers are adapted to operate in a burst mode over a range between a light load and no load.
 42. The low input-current-harmonic boost rectifier of claim 23, wherein the inductive elements are magnetically coupled.
 43. The low input-current-harmonic boost rectifier of claim 23, wherein one or more inrush current control circuits are coupled between the input voltage and the capacitors.
 44. The low input-current-harmonic boost rectifier of claim 23, wherein the input stage further comprises input electromagnetic interference (EMI) filters.
 45. The low input-current-harmonic boost rectifier of claim 23, wherein the input bridge diodes are synchronous rectifiers.
 46. The low input-current-harmonic boost rectifier of claim 23, wherein each switch comprises a MOSFET or an IGBT with an antiparallel diode.
 47. The low input-current-harmonic boost rectifier of claim 23 wherein a plurality of auxiliary series switches are coupled to said switches through said at least one transformer, wherein the controller is further adapted to operate said auxiliary switches according to at least one of phase-shift or pulse width modulation.
 48. The low input-current-harmonic boost rectifier of claim 47 wherein said auxiliary switches are coupled to said switches through said at least one transformer to form at least one full-bridge converter.
 49. The low input-current-harmonic boost rectifier of claim 47, wherein the transformer comprises a primary winding coupled between the junction of the auxiliary switches and the neutral node.
 50. The low input-current-harmonic boost rectifier of claim 47, wherein the transformer comprises a primary winding coupled through one or more blocking capacitors to the neutral node. 